当从壁装电源插座操作开关转换器时，部分吸收的能量为负载供电，而其余部分则会损失并散热。
为了了解给定应用中的损耗量 - 假设一个机顶盒适配器 - 只需在操作时将手放在盖子上并感受到热量：温暖的外壳意味着在冷的情况下显着损失（没有双关语意图）
）表征高效电源。
如果您现在按下待机按钮，您希望盒子在几分钟后冷却下来。
但是，在某些情况下，您仍然可以感受到温暖的外壳，告诉您尽管待机仍然存在损失。
开关转换器中的损耗已成为众多出版物的主题。
在这篇简短的论文中，我们将只关注典型转换器的图1所示的次级侧。
图1 - 在今天的大多数转换器中，控制位于次级侧。
这里有一个TL431，它承载一个复合误差放大器和一个参考电压。
在此应用电路中，您可以识别TL431，其参考引脚接收您要监视的变量的一部分Vout。
TL431阴极电流I1相对于负载电流进行调节，以保持恒定的输出电压。
目标是通过光耦合器光学传输的电流修改图1左侧所示的初级侧反馈电压。
该反馈电压将最终在电流模式转换器中设置转换器峰值电流。
光耦合器受定义为的电流传输比（CTR）的影响
（1）
其中IC是集电极电流，IF是LED正向电流。
TL431集成了11个双极晶体管，形成运算放大器和2.5V基准电压。
该部件直接由阴极吸收的电流操作。
该电流必须至少为1 mA，如元件数据手册中所述。
这1毫安是在初级侧产生IFB所需的反馈电流之上。
利用LED 1-V正向压降，电阻器Rbias利用光耦LED构建了一个廉价的电流发生器。
如果Rbias为1kΩ，则大致注入1 mA。
假设初级侧上拉电阻RFB为20kΩ，内部Vdd为5 V，则反馈电压在5 V和0.3 V之间移动所需的最大电流（光耦合器饱和电压）被发现为
（2）
考虑到25％的CTR，该集电极电流转换为LED正向电流
（3）
添加RFB提供的1 mA额外偏置电流，总输出电流为1.94 mA。
如果此输出为20 V（笔记本适配器的典型值），则会永久消散
（4）
与典型适配器（45至65 W）提供的功率相比，它可能不被视为大量浪费。
但是，当您努力将空载功率限制在100 mW以下时，您可以追踪每一个可以节省的功率。
在（4）中，部分结果是RFB带来的1mA偏置电流。
这个电流对于确保在全功率（高直流增益）下正确的TL431偏置非常重要，但在空载待机状态下，我们可以想出一种方法来消除它。
图2显示了安森美半导体的专有电路，该电路可以很好地偏置器件，但是当转换器开始在轻负载下工作时，偏置逐渐降低，在待机状态下完全消失。
图2 - 简单的峰值整流器有助于在空载情况下消除偏置电流。
围绕D1，C1和Rbias构建的峰值检测器确保在高功率下的完全偏置，其中C1两端的电压等于Vout。
当控制器进入跳跃周期模式时，C1上的峰值电压保持不变，但谷值电压因C1Rbias时间常数而下降。
当进入深度待机模式时，Rbias中的电流减小到完全消失：从（4）的总数中节省20 mW。
点击率和次要侧功率
如（3）所示，初级侧电流通过光耦合器CTR反射到次级侧。
不幸的是，为了减少待机模式下的损耗，反馈电阻RFB通常保持高电平（20-40kΩ）以强制产生低集电极电流。
在此模式下，光耦合器的点击率较低，通常低至25％。
低CTR的缺点被视为次级侧的额外耗散功率，尤其是在如（4）所示的高Vout时。
在这种情况下，为什么不选择更高的CTR光电耦合器呢？
诸如SFH615之类的流行参考具有不同的CTR等级。
例如，-2表示在10mA电流下CTR从63％变化到125％。
如果选择-4，则CTR现在在160％和320％之间变化。
即使在低反馈电流下，您也不会获得160％的点击率，但LED电流可能明显低于-2版本。
使用-4测量电路板时，CTR为43％，而-2为25％。
考虑到TL431偏置电流在待机状态下消失，次级侧功率下降
（5）
这是（4）带来的四分之一的力量。
可以选择更高的CTR，可能的选择是FOD817S即使在低集电极电流下也能达到75％的CTR。
在这种情况下，LED偏压降至313μA，导致功率损失
（6）
高CTR光电耦合器的选择并非微不足道。
光耦合器受寄生元件的影响，特别是在集电极和发射极之间看到的电容器。
该电容与RFB一起产生一个低频极点，如果交叉较高，可能会影响转换器的相位裕量。
重要的是要描述这个极点并量化其对稳定性的影响。
它的位置与CTR等级有关。
如果您需要高带宽，请选择低CTR光电耦合器。
待机功率将受到影响，但杆位置应该不是问题。
相反，高CTR光电耦合器适用于低功耗系统，但您可能必须选择低交叉频率，以便极点不会妨碍稳定性。
在任何情况下，如果您决定用更高CTR类型更换光耦合器，请务必再次测量转换器的动态响应以检查是否存在任何可能的故障。
以下数字显示了升级光耦合器CTR带来的空载待机功耗性能改善。
该板采用新型低压NCP1256控制器，包括欠压检测。
所有测量均在230 V rms输入电压下进行。
存在解决方案以在无负载条件下操作时降低转换器待机功率。
增加分压器检测网络值是一个明显的途径，但降低TL431偏置电流或改善光耦合器CTR很容易实现奖励选项。
以上来自于谷歌翻译
以下为原文
When operating a switching converter from the wall outlet, part of the absorbed energy feeds the load while the rest is lost and dissipated in heat. To get an idea of the loss amount in a given application – assume a set-top box adapter – simply place your hand on the cover while it operates and feel the heat: a wARM enclosure implies significant losses while a cold case (no pun intended) characterizes a highly-efficient power supply. If you now press the standby button, you expect the box to cool down after several minutes. However, in some cases, you can still feel a warm enclosure, telling you that losses still exist despite standby.
Losses in switching converters have been the subject of numerous publications. In this short paper, we will only concentrate on the secondary side shown in Figure 1 for a typical converter.
![]()
Figure 1 - In most of today’s converters, control is located in the secondary side. Here a TL431 hosting a compound error amp and a reference voltage.
In this application circuit, you recognize a TL431 whose reference pin receives a portion of the variable you want to monitor, Vout. The TL431 cathode current I1 is adjusted in relationship to the load current so as to maintain a constant output voltage. The goal is to modify the primary-side feedback voltage shown in Figure 1’s left side via a current optically transmitted by the optocoupler. This feedback voltage will ultimately set the converter peak current in a current-mode converter. The optocoupler is affected by a current transfer ratio (CTR) defined as
![]() (1)
where IC is the collector current and IF the LED forward current.
A TL431 aggregates 11 bipolar transistors to form an op amp and a 2.5-V reference voltage. The part directly operates from the current absorbed by the cathode. This current must be 1 mA minimum as stated in the component data-sheet. This 1 milliamp comes on top of the feedback current necessary to produce IFB in the primary side. Capitalizing on the LED 1-V forward drop, resistor Rbias builds a cheap current generator with the optocoupler LED. If Rbias is 1 kΩ, then you roughly inject 1 mA. Assuming a primary-side pull-up resistor RFB of 20 kΩ and an internal Vdd of 5 V, the maximum current needed to move the feedback voltage between 5 V and 0.3 V (the optocoupler saturation voltage) is found to be
![]() (2)
Considering a 25% CTR, this collector current translates into a LED forward current of
![]() (3)
Adding the 1-mA extra bias current provided by RFB, you total a current of 1.94 mA, drawn from the output. If this output is 20 V (typical value for a notebook adapter), you permanently dissipate
![]() (4)
Compared to the power delivered by a typical adapter (45 to 65 W), it may not be regarded as a tremendous amount of waste. However, when you struggle to limit the no-load power below the 100-mW barrier, you chase every mW you can save. In (4), part of the result is the 1-mA bias current brought by RFB. This current is important to ensure a proper TL431 bias at full power (high dc gain) but in no-load standby, we could think of a way to remove it.
Figure 2 shows you an ON Semiconductor proprietary circuit that nicely biases the part at high power but as the converter starts operating in light load, the bias gradually decreases to completely disappear in standby.
![]()
Figure 2 - A simple peak rectifier helps removing the bias current in a no-load situation.
The peak detector built around D1, C1 and Rbias ensures a full bias at high power where the voltage across C1 equals Vout. When the controller enters skip cycle mode, the peak voltage on C1 remains the same but the valley voltage drops owing to the C1Rbias time constant. The current in Rbias reduces to fully disappear when a deep standby mode is entered: you save 20 mW from (4)’s total.
CTR and Secondary-Side Power
As indicated by (3), the primary-side current reflects to the secondary via the optocoupler CTR. Unfortunately, to reduce losses in standby mode, the feedback resistor RFB is usually kept high (20-40 kΩ) to force a low collector current. In this mode, the optocoupler exhibits a poor CTR, often as low as 25%. The drawback of a low CTR is seen as extra dissipated power in the secondary side, especially at a high Vout as indicated by (4). In this case, why not select higher CTR optocouplers? A popular reference such as SFH615 comes in different CTR grades. For instance, the -2 shows a CTR varying from 63 to 125% at a 10-mA current. If you select a -4, the CTR now varies between 160% and 320%. Even if at a low feedback current you won’t get a 160% CTR, it is likely that the LED current is significantly lower than the -2 version. Measurement on the board with -4, shows a CTR of 43% versus 25% with the -2. Considering the disappearance of the TL431 bias current in standby, the secondary-side power drops
![]() (5)
which is a quarter of the power brought by (4). Higher CTRs can be selected and a possible option is the FOD817S that exhibits a CTR of 75% even at low collector currents. In this case, the LED bias drops to 313 µA, inducing a power loss of
![]() (6)
The selection of high-CTR optocouplers is not trifling. An optocoupler is affected by parasitic elements, in particular a capacitor seen between collector and emitter. This capacitor together with RFB creates a low-frequency pole that can affect the converter phase margin if crossover is high. It is important to characterize this pole and quantify its impact on stability. Its position varies in relationship to the CTR grade. If you want high-bandwidth, select low-CTR optocouplers. Standby power will suffer but the pole position should be less of a problem. On the opposite, high-CTR optocouplers are good for low-power systems but you may have to select a low crossover frequency so that the pole does not hamper stability. In any case, if you decide to replace your optocoupler by a higher-CTR type, make sure to measure the converter’s dynamic response again to check for any possible troubles.
The below numbers show the no-load standby power performance improvement brought by upgrading the optocoupler CTR. The board uses the new low-voltage NCP1256 controller which includes brown-out sensing. All measurements are carried at a 230-V rms input voltage.
![]()
Solutions exist to lower a converter standby power when operated in a no-load condition. Increasing the voltage divider sensing network value is one obvious path but reducing the TL431 bias current or improving the optocoupler CTR are simple to implement rewarding options.
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